Conventionally, a motor used in an electric power steering apparatus is, in general, a permanent magnet synchronous motor (PMSM), which is driven by a three-phase sinusoidal current. As a control system for driving the motor, a control system called vector control is widely used. However, since there is a strong demand for a reduction in size of the electric power steering apparatus, a brushless DC motor tends to be used as a motor suitable for the reduction in size.
Under such circumstances, a motor drive control device using the advance angle control system for the conventional motor for the electric power steering apparatus will be explained with reference to FIG. 1.
In a structure of the motor drive control device, a main path leading to the motor 1 is connected to the back of a current command value calculating unit 100 that controls an electric current of a motor 1 via subtracters 20-1, 20-2, and 20-3 that detect errors between phase current command values Iavref, Ibvref, and Icvref and motor currents Ia, Ib, and Ic, a PI control unit 21 that inputs respective error signals from the subtracters 20-1, 20-2, and 20-3, a PWM control unit 30 that inputs voltages va, vb, and vc from the PI control unit 21, and an inverter 31 that converts a direct current into an alternating current. Current detecting circuits b 32-1, 32-2, and 32-3, which detect the motor currents Ia, Ib, and Ic, are arranged between the inverter 31 and the motor 1. A feedback control system B, in which the detected motor currents Ia, Ib, and Ic are fed back to the subtracters 20-1, 20-2, and 20-3, respectively, is formed.
Next, the current command value calculating unit 100 will be explained. First, concerning inputs thereof, a torque command value Tref calculated from a torque detected by a not-shown torque sensor, a rotation angle θe of a rotor in the motor 1 detected by a position detecting sensor 11 connected to the motor 1, and an electrical angular velocity we calculated by a differentiating circuit 24 are inputted. A converting unit 101 calculates counter-electromotive forces ea, eb, and ec with the electrical angular velocity we and the rotation angle θe of the rotor as inputs. Next, a three-phase/two-phase converting unit 102 converts the counter-electromotive forces ea, eb, and ec into a d-axis component voltage ed and a q-axis component voltage eq. A q-axis command current calculating unit 108 calculates a current command value on a q-axis Iqref with the d-axis component voltage ed and the q-axis component voltage eq as inputs. However, in this case, a current command value on a d-axis Idref is calculated as 0. In other words, in the following output equation of a motor,Tref×ωm=3/2(ed×Id+eq×Iq)  (1)when Id =Idref=0 is inputted, the equation is calculated as follows.Iq=Iqref=2/3(Tref×ωm/eq)  (2)Phase current command values Iavref, Ibvref, and Icvref are calculated on the basis of a current command value Iqref from the q-axis command current calculating unit 108 and an advance angle Φ of advance angle control described later. In other words, a two-phase/three-phase converting unit 109 calculates the phase current command values Iavref, Ibvref, and Icvref based on the advance angle Φ calculated in the advance angle calculating unit 107 and the current command value Iqref calculated in the q-axis command current calculating unit 108.
Note that a function such as Φ=a cos(ωb/ωm) or Φ=K(1−(ωb/ωm)) is used empirically (“a cos” represents cos1). In addition, a motor base angular velocity ωb is a motor limit angular velocity at the time when the motor 1 is driven without using field-weakening control. FIG. 2 shows a relation between a torque T and a motor speed n (the angular velocity ωe) and shows an example of the limit angular velocity ωb in the case in which there is no field-weakening control.
Next, the advance angle control will be explained.
While the motor 1 is not rotating at high speed, that is, while a mechanical angular velocity ωm of the motor 1 is lower than the motor base angular velocity ωb, it is possible to output a torque complying with the torque command value Tref if the phase current command values Iavref, Ibvref, and Icvref in accordance with a value calculated from the current command value Iqref by the two-phase/three-phase converting unit 109 regardless of the advance angle Φ. This means that, as the electric power steering device, wheel operation by a driver is executed smoothly.
However, when the motor 1 rotates at high speed, that is, the mechanical angular velocity ωm of the motor is higher than the motor base angular velocity ωb, an angular velocity higher than the base angular velocity ωb cannot be realized unless control taking into account the advance angle Φ is executed. When this high-speed rotation of the motor 1 is considered from a viewpoint of the electric power steering apparatus, in the case of sudden steering of a wheel for turn in parking a car or emergency shelter, steering feeling of the driver is deteriorated because the motor 1 does not follow the wheel operation.
There is a control system called field-weakening control as torque control at the time of high-speed rotation of a motor. There is advance angle control as a specific method of the field-weakening control. Details of this advance angle control system are described in U.S. Pat. No. 5,677,605 (C1) and C. C. Chan et al “Novel Permanent Magnet Motor Drivers for Electric Vehicles” IEEE Transaction on Industrial electronics (Vol 43, No. 2 Apr. 1996, page 335, FIG. 5). A characteristic of the advance angle control system is to advance a phase of the current command value Iqref by the angle Φto create a field-weakening component. In FIG. 10(B), when the current command value Iqref is advanced by the angle Φ, Iqref×sin Φ is generated as a d-axis component and Iqref×cos Φ is generated as a q-axis component. Here, Iqref×sin Φ acts as a field-weakening component and Iqref×cos Φ acts as a torque component.
As a motor drive control system used in the electric power steering apparatus, vector control, which is adapted to generate a rotating magnetic field from a control device via an inverter on the basis of rotating position of a rotor to control to drive rotation of the rotor, is adopted. In other words, the vector control is adapted to, in plural exciting coils arranged at intervals of a predetermined angle on an outer peripheral surface of the rotor, control rotation drive for the rotor by sequentially switching excitation of the respective exciting coils using a control circuit according to a rotor position.
This type of vector control is disclosed in, for example, JP-A-2001-18822. FIG. 3 is a block diagram showing an example of drive control for a motor 56 according to the vector control.
In FIG. 3, a main path of a command signal leading to the motor 56 from a command current determining unit 51, which determines a control command value of the motor 56, via a PI control unit 52, a two-phase/three-phase coordinate converting unit 53, a PWM voltage generating unit 54, and an inverter 55 is formed. Current sensors 571 and 572 are arranged between the inverter 55 and the motor 56. A feedback path, in which a three-phase/two-phase coordinate converting unit 59 converts a motor current detected by the current sensors 571 and 572 into a two-phase current to feed back two-phase current components Iq and Id to subtracting circuits 581 and 582 arranged between the command current determining unit 51 and the PI control unit 52, is formed.
With this control system, the command current determining unit 51 receives the torque command value Tref calculated from a torque detected by the torque sensor and a rotor rotating angle θ and an electrical angle ω detected by the position detecting sensor to determine current command values Idref and Iqref. These current command values Idref and Iqref are subjected to feedback correction by the two-phase current components Iq and Id, which are converted into two phases by the three-phase/two-phase coordinate converting unit 59 in the feedback pass, in the subtracting circuits 581 and 582, respectively. In other words, the subtracting circuits 581 and 582 calculate errors between the two-phase current components Id and Iq and the current command values Idref and Iqref. Thereafter, PI control units 521 and 522 calculate signals, which indicate duty of PWM control, as command values Vd and Vq in forms of a d component and a q component. The two-phase/three-phase coordinate converting unit 53 inversely converts the d component and the q component into three-phase components Va, Vb, and Vc. The inverter 55 is subjected to the PWM control on the basis of the three-phase command values Va, Vb, and Vc and an inverter current is supplied to the motor 56 to control rotation of the motor 56.
Note that reference numeral 61 denotes a vehicle speed sensor; 62, a sensitive are a judging circuit; 63, a coefficient generating circuit; 64, a basic assist force calculating circuit; 65, a return force calculating circuit; 66, an electrical angle converting unit; 67, an angular velocity converting unit; and 68, a non-interference control correction value calculating unit.
In the case of the vector control described above, the current command values Idref and Iqref are determined on the basis of the torque command value Tref, the electrical angle ω, and the rotation angle θ. Feedback currents Iu and Iw of the motor 56 are converted into three-phase currents Iu, Iv, and Iw and, then, converted into two-phase current components Id and Iq. Thereafter, the subtracting circuits 582 and 581 calculate errors between the two-phase current components Id and Iq and the current command values Idref and Iqref. Current control by the PI control is executed according to the errors, whereby command values Vd and Vq to the inverter 55 are calculated. Then, the two-phase/three-phase coordinate converting unit 53 inversely converts the command values Vd and Vq into the three-phase command values Va, Vb, and Vc, whereby the inverter 55 is controlled to perform drive control for the motor 56.
Incidentally, the d-axis component and the q-axis component generated by the advance angle control simply advance the current command value Iqref by the phase Φ. Thus, Iqref×sin Φ on the d-axis and Iqref×cos Φ on the q-axis are restricted to a fixed relation and a quantitative balance is not always optimized. As a result, a motor terminal voltage is saturated at the time of high-speed rotation and a motor current cannot follow a current command value, whereby torque ripple increases and motor noise also increases. Therefore, as the electric power steering apparatus, inconveniences are caused in that, for example, a driver feels abnormal vibration through a wheel at the time of rapid wheel steering and motor noise is cased to give unpleasant feeling to the driver.
In the case of the vector control described above, a detection current of the motor 56 and an output of the inverter 55 are in three phases and the feedback control system is in two phases. It is necessary to control to drive the motor 56 by inversely converting two phases into three phases in the two-phase/three-phase coordinate converting unit 53 in this way. Thus, there is a problem in that the entire control system is complicated because the two-phase/three-phase conversion and the three-phase/two-phase conversion are mixed.
In the control of the motor 56, if it is possible to maintain linearity of the control system, control responsiveness is improved. Thus, the control is easy and a control target is easily attained. However, various nonlinear factors are included in drive control for the motor 56. As a factor causing nonlinearity of motor drive, for example, there is dead time of inverter control. Although an FET is used as a switching element of an inverter, the FET is not an ideal switching element. In order to prevent short circuit in upper and lower arms, a period for setting both FETs of the upper and lower arms in an off state (dead time) is provided. A nonlinear element of a switching transition state is included in a motor current generated by switching of the FETs having such dead time. In addition, a nonlinear element is also included in a detection element, a detecting circuit, and the like for detecting a motor current.
As a result, for example, a nonlinear element generated in an a-phase current Ia is included in the d-axis current component Id and the q-axis current component Iq by the d-q conversion in the three-phase/two-phase coordinate converting unit 59 of the feedback system. Therefore, current control is performed on the basis of the current components Id and Iq, the command values Vd and Vq from the PI control units 522 and 521 to the inverter 55 are calculated, a d-phase and a q-phase are inversely converted into an a-phase, a b-phase, and a c-phase in the two-phase/three-phase coordinate converting unit 53, and three-phase command values Va, Vb, and Vc are calculated. Consequently, the nonlinear element originally included in the a-phase current Ia is diffused to the command values Va, Vb, and Vc of the inverter 55 by the d-q conversion. Thus, the nonlinear element is included in the command values of not only the a-phase but also the b-phase and the c-phase. In other words, in the case of the conventional control system, despite the fact that the motor is driven in three phases, feedback current control is calculated in two phases and the command values Vd and Vq determined in two phases are converted into the three-phase command values Va, Vb, and Vc formally and controlled. Thus, the nonlinear element is diffused.
Therefore, according to the conventional motor control, there is a problem in that torque ripple is large and noise of the motor is also large. When such motor control is applied to the electric power steering apparatus, the electric power steering apparatus cannot assist steering accurately and smoothly following wheel operation. Thus, there is a problem in that a driver feels vibration at the time of steering and noise increases.
The invention has been devised because of the circumstances described above and it is an object of the invention to provide a motor and a drive control device for the motor, in which torque ripple is reduced and noise is reduced by controlling nonlinear elements included in motor control in a state in which the nonlinear elements are separated into respective phases, and also provides an electric power steering apparatus that adopts the motor and the drive control device to have an improved steering performance and satisfactory steering feeling.
It is another object of the invention to provide a motor drive control device, in which a motor terminal voltage is not saturated even at the time of high-speed rotation of a motor, torque ripple is reduced and motor noise is reduced, and an electric power steering apparatus in which noise is reduced at the time of rapid steering of a wheel and with which wheel operation can follow the steering smoothly.